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A forum for the exchange of circuits, systems, and software for real-world signal processing
Li-Ion BATTERY CHARGING REQUIRES ACCURATE VOLTAGE SENSING (page 3)
Quad-SHARC in CQFP—A 480-MFLOPS DSP Powerhouse (page 10)
Ask the Applications Engineer—Capacitive Loads on Op Amps (page 19)
Complete contents on page 3
a
Volume 31, Number 2, 1997
Editor’s Notes
NEW FELLOW
We are pleased to note that Woody
Beckford was introduced as the
newest Fellow at our 1997 General
Technical Conference.
Fellow
, at
Analog Devices, represents the
highest level of achievement that a
technical contributor can achieve,
on a par with Vice President. The
criteria for promotion to Fellow are
very demanding. Fellows will have
earned universal respect and recognition from the technical
community for unusual talent and identifiable innovation at the
state of the art; their creative technical contributions in product
or process technology will have led to commercial success with a
major impact on the company’s net revenues.
Their attributes include roles as mentor, consultant, entrepreneur,
organizational bridge, teacher, and ambassador. They must also
be effective leaders and members of teams and in perceiving
customer needs. Woody’s technical abilities, accomplishments, and
personal qualities well-qualify him to join Fellows Derek Bowers
(1991), Paul Brokaw (1980), Lew Counts (1984), Barrie Gilbert
(1980) Jody Lapham (1988), Fred Mapplebeck (1989), Jack
Memishian (1980), Doug Mercer (1995), Mohammad Nasser
(1993), Wyn Palmer (1991), Richie Payne (1994), Carl Roberts
(1992), Paul Ruggerio (1994), Brad Scharf (1993), Mike Timko
(1982), Bob Tsang and Mike Tuthill (1988), Jim Wilson (1993),
and Scott Wurcer (1996).
WOODROW BECKFORD
A manufacturer of high-
performance analog and mixed-
signal products must be able
rapidly and economically to test
products meeting state-of-the-art
specs in production quantities for
market success and profitability.
ADI’s Component Test Systems
division specializes in designing
and manufacturing advanced test
systems to production-test our leadership products at low cost.
Woody Beckford, the father of the latest series, the CTS-5000s,
conceived the entire system architecture—hardware and software,
designed the initial electrical and mechanical hardware, and
directed the software development. The result, VXI-based testing
with autocalibration (no trimming pots) and a loaded cost 1/3 the
cost of anything comparable. But nothing comparable exists; many
CTS-5000 capabilities, unavailable at any price, have been key to
ADI’s continued leadership in high-speed converters.
Woody was born and raised in Massachusetts and graduated from
Northeastern University in 1982 with a BS in Physics. His co-op
program jobs were at the MIT Bates Linear Accelerator (high-
energy physics) and LTX Corporation (ATE). After graduation,
he came to work for ADI’s CTS division. He has worked on just
about every generation of CTS test equipment—the 2000, 3000,
and 5000 series. In his spare time he enjoys amateur radio
(N1IBY), listening to music, riding motorcycles, and target
shooting. He is married, with 3 children.
b
Dan.Sheingold@analog.com
THE AUTHORS
Joe Buxton
(page 3) is a Senior
Design Engineer in Santa Clara,
CA, for ADI’s Power Management
product line. He is currently
designing battery charger and
switching regulator ICs. Previously
he was an Applications Engineer,
writing numerous articles and
developing SPICE macromodels for
ADI components. Joe has a BSEE
(1988) from the University of California, Berkeley. In his leisure
time, he enjoys running, skiing, reading, and travel.
Mike Walsh
(page 5) is a Senior
Product Engineer in the High-
Speed Converter group, in
Wilmington, MA, working on high-
resolution ADCs and mixed-signal
consumer video ASICs. He joined
Analog Devices in 1989, after
graduating from Boston University
with a MSEE. In his spare time,
Mike enjoys woodworking and
playing with his two daughters.
Bob Scannell
(page 10) is a
Product Marketing Manager in the
Multichip Products group of ADI,
in Greensboro, NC. Bob has been
involved with research, design, and
marketing of DSP multiprocessors
and multichip modules for 12
years. He holds a MS Computer
Engineering degree from USC and
a BSEE from UCLA. When away
from work, Bob enjoys woodworking and travel.
Grayson King
(page 19), an
Applications Engineer in the Central
Applications group in Wilmington,
MA, has a BSEE from Clarkson
University. In addition to providing
customer support for linear and
converter products, Grayson is
currently working to develop
computer tools to aid designers in
product selection. He also enjoys
telemark skiing, white-water kayaking, and finding new ways to
make a simple audio amplifier.
[
More authors on page 22
]
Cover:
The cover illustration was designed and executed by
Shelley Miles
, of
Design Encounters
,
Hingham MA.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106
Published by Analog Devices, Inc. and available at no charge to engineers and
scientists who use or think about I.C. or discrete analog, conversion, data handling
and DSP circuits and systems. Correspondence is welcome and should be addressed
to Editor,
Analog Dialogue
, at the above address. Analog Devices, Inc., has
representatives, sales offices, and distributors throughout the world. Our web site is
http://www.analog.com/
. For information regarding our products and their
applications, you are invited to use the enclosed reply card, write to the above address,
or phone 617-937-1428, 1-800-262-5643 (U.S.A. only) or fax 617-821-4273.
2
ISSN 0161–3626
©Analog Devices, Inc. 1997
Analog Dialogue 31-2 (1997)
Li-Ion Battery
Charging Requires
Accurate Voltage
Sensing
New Battery Charger Controller
Guarantees 61% Final Battery
Voltage Accuracy
by Joe Buxton
Lithium-Ion (Li-Ion) batteries are gaining popularity for portable
systems due to their increased capacity at the same size and weight
as the older NiCad and NiMH chemistries. For example, a portable
computer equipped with a Li-Ion battery can have a longer
operating time than a similar computer equipped with a NiMH
battery. However, designing a system for Li-Ion batteries requires
special attention to the charging circuitry to ensure fast, safe, and
complete charging of the battery.
A new battery-charging IC, the ADP3810*, is designed specifically
for controlling the charge of 1-to-4-cell Li-Ion batteries. Four high-
precision fixed final battery-voltage options (4.2˚ V, 8.4˚ V, 12.6˚ V,
and 16.8˚ V) are available; they guarantee the
The ADP3810 and ADP3811:
Figure 1 shows the functional
diagram for the ADP3810/3811 in a simplified CCCV charger
circuit. Two “
g
m
” amplifiers (voltage input, current output) are
key to the IC’s performance. GM1 senses and controls the charge
current
via shunt resistance, R
CS
, and GM2 senses and controls
the final battery
voltage
. Their outputs are connected in an analog
“OR” configuration, and both are designed such that their outputs
can only pull up the common COMP node. Thus, either the current
amplifier or the voltage amplifier is in control of the charging loop
at any given time. The COMP node is buffered by a “
g
m
” output
stage (GM3), the output current of which directly drives the dc-
dc converter control input (via an opto-coupler in isolated
applications).
V
BAT
OUT
DC/DC
CONVERTER
I
CHAR
GE
V
IN
IN
CTRL
BATTERY
V
RCS
R
CS
RETURN
GND
2.0V
0.
1
m
F
ADP3811
ONLY
0.
1
m
F
R1
R3
R2
V
CS
V
CC
V
REF
V
S
ENS
E
V
REF
UVLO
ADP3810
ONLY
R1
1.
5M
V
8
0k
V
UVL
O
V
CTRL
R2
GM1
BUFFER
V
REF
GM 2
UVLO
200
V
I
O
U
T
ADP3810/
ADP3811
OUT
100
m
A
1.2V
GM 3
1% final battery
voltage specification that is so important in charging Li-Ion
batteries. A companion device, the ADP3811, is similar to the
ADP3810, but its final battery voltage is user-programmable to
accommodate other battery types. Both ICs accurately control the
charging current to realize fast charging at currents of 1 ampere
or more. In addition, they both have a precision 2.0-V reference,
and a direct opto-coupler drive output for isolated applications.
Li-Ion Charging:
Li-Ion batteries commonly require a constant
current, constant voltage (CCCV) type of charging algorithm. In
other words, a Li-Ion battery should be charged at a set current
level (typically from 1 to 1.5 amperes) until it reaches its final
voltage. At this point, the charger circuitry should switch over to
constant voltage mode, and provide the current necessary to hold
the battery at this final voltage (typically 4.2˚ V per cell). Thus, the
charger must be capable of providing stable control loops for
maintaining either current or voltage at a constant value, depending
on the state of the battery.
The main challenge in charging a Li-Ion battery is to realize the
battery’s full capacity without overcharging it, which could result
in catastrophic failure. There is little room for error, only
±
COMP
GND
C
C
R
C
Figure 1. Block diagram of the ADP3810/3811 in a simpli-
fied battery-charging circuit.
The ADP3810 includes precision thin-film resistors to divide down
the battery voltage accurately and compare it to an internal 2.0-V
reference. The ADP3811 does not include these resistors, so the
designer can program any final battery voltage with an external
resistor pair according to the formula below. A buffer amplifier
provides a high-impedance input to program the charge current
using the VCTRL input, and an under voltage lock-out (UVLO)
circuit ensures a smooth start-up.
æ
ç
ö
÷
R
1
R
2
V
BAT
=
2.000 V
´
1
+
IN THIS ISSUE
Volume 31, Number 2, 1997, 24 Pages
Editor’s Notes (New Fellow: Woody Beckford), Authors, . . . . . . . . . . . . . ˚ 2
Li-Ion battery charging requires accurate voltage sensing . . . . . . . . . . . . . . . .
˚ 3
Pin-compatible 14-bit monolithic ADCs: First to sample from 1-10 MSPS
(AD924x) 5
200-MHz 16˚
´
˚ 16 video crosspoint switch IC
(AD8116)
. . . . . . . . . . . . . . .
˚ 6
Selecting mixed-signal components for digital communications systems (IV) . . .
˚ 7
Quad-SHARC DSP in CQFP—a 480-MFLOPS powerhouse
(AD14060) . 10
Digital signal processing 101—an introductory course in DSP system design—II . .
11
New-Product Briefs:
ADCs and DACs, R-DAC, Audio Playback . . . . . . . . . . . . . . . . . . . . 15
Amplifiers, Mux, Reference, DC-DC . . . . . . . . . . . . . . . . . . . . . . . . . 16
Power Management, Supervisory Circuits . . . . . . . . . . . . . . . . . . . . . 17
Temp Sensor, Codec, Communications & ATE ICs . . . . . . . . . . . . . . 18
Ask The Applications Engineer—25:
Op amps driving capacitive loads . . .
19
Worth Reading, More authors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Potpourri . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
1%.
Overcharging by more than +1% could result in battery failure,
but undercharging by more than 1% results in reduced capacity.
For example, undercharging a Li-Ion battery by only 100˚ mV
(–2.4% for a 4.2-V Li-Ion cell) results in about a 10% loss in
capacity. Since the room for error is so small, high accuracy is
required of the charging-control circuitry. To achieve this accuracy,
the controller must have a precision voltage reference, a low-offset
high-gain feedback amplifier, and an accurately matched resistance
divider. The combined errors of all these components must result
in an overall error less than
±
1%. The ADP3810, combining these
elements, guarantees the overall accuracy of ± 1%, making it an
excellent choice for Li-Ion charging.
±
Analog Dialogue 31-2 (1997)
3
To understand the “OR” configuration, assume that a fully
discharged battery is inserted in the charger. The voltage of the
battery is well below the final charge voltage, so the VSENSE input
of GM2 (connected to the battery) brings the positive input of
GM2 well below the internal 2.0-V reference. In this case, GM2
wants to pull the COMP node low, but it can only pull up, so it
has no effect at the COMP node. Since the battery is dead, the
charger starts to increase the charge current and the current loop
takes control. The charge current develops a negative voltage across
the 0.25-W current-shunt resistor (RCS). This voltage is sensed
by GM1 through the 20-k
specifications of the charger are controlled by the ADP3810/3811,
which guarantees the final voltage within
1%.
The current drive of the ADP3810/3811’s control output directly
connects to the photo-diode of an opto-coupler with no additional
circuitry. Its 4-mA output current capability can drive a variety of
opto-couplers—an MOC8103 is used here. The current of the
photo-transistor flows through R
F
, setting the voltage at the 3845’s
COMP pin and thus controlling the PWM duty cycle. The
controlled switching regulator is designed so that increased LED
current from the opto-coupler reduces the duty cycle of the
converter.
While the signal from the ADP3810/3811 controls the
average
charge current, the primary side should have a cycle by cycle limit
of the switching current. This current limit has to be designed
such that, with a failed or malfunctioning secondary circuit or
opto-coupler, or during start up, the primary power circuit
components (the FET and transformer) won’t be over-stressed.
When the secondary side V
CC
rises above 2.7˚ V, the ADP3810/
3811 takes over and controls the average current. The primary
side current limit is set by the 1.6-W current sense resistor
connected between the power NMOS transistor, IRFBC30, and
ground.
The ADP3810/3811, the core of the secondary side, sets the overall
accuracy of the charger. Only a single diode is needed for
rectification (MURD320) and no filter inductor is required. The
diode also prevents the battery from back driving the charger when
input power is disconnected. A 1000-
±
resistor (R3). At equilibrium,
(
I
CHARGE
R
CS
)/
R
3
˚ =˚ –
V
CTRL
/80˚ k
W
W
. Thus the charge current is
maintained at
R
3
R
CS
If the charge current tends to exceed the programmed level, the
V
CS
input of GM1 is forced negative, which drives the output of
GM1 high. This in turn pulls up the COMP node, increasing the
current from the output stage, reducing the drive of the dc/dc
converter block (which could be implemented with various
topologies such as a flyback, buck, or linear stage), and finally,
reducing the charge current. This negative feedback completes the
charge current control loop.
As the battery approaches its final voltage, the inputs of GM2
come into balance. Now GM2 pulls the COMP node high and
the output current increases, causing the charge current to
decrease, maintaining
V
SENSE
and
V
REF
equal. Control of the
charging loop has changed from GM1 to GM2. Because the gain
of the two amplifiers is very high, the transition region from current
to voltage control is very sharp, as Figure 2 shows. This data was
measured on a 10-V version of the off-line charger of Figure 3.
Complete Off-Line Li-Ion Charger:
Figure 3 shows a complete
charging system using the ADP3810/3811. This off-line charger
uses the classic flyback architecture to create a compact, low cost
design. The three main sections of this circuit are the primary-
side controller, the power FET and flyback transformer, and the
secondary-side controller. This design uses an ADP3810, directly
connected to the battery, to charge a 2-cell Li-Ion battery to 8.4˚ V
at a programmable charge current from 0.1 to 1˚ A. The input range
is from 70˚ to˚ 220˚ V˚ ac—for universal operation. The primary side
pulsewidth modulator used here is the industry-standard 3845,
but other PWM components could be used. The actual output
V
CTRL
80 k W
I
CHARGE
=
F capacitor (CF1) maintains
stability when no battery is present. R
CS
senses the average current
(see above), and the ADP3810 is connected directly (or ADP3811
through a divider) to the battery to sense and control its voltage.
With this circuit, a complete off-line Li-Ion battery charger is
realized. The flyback topology combines an AC/DC converter with
the charger circuitry to give a compact, low-cost design. The
accuracy of this system depends on the secondary side controller,
the ADP3810/3811. The device’s architecture also works well in
other battery charging circuits. For example, a standard dc-dc buck
type of charger can easily be designed by pairing the an ADP3810
and an ADP1148. A simple linear charger can also be designed
with just the ADP3810 and an external pass transistor. In all cases,
the inherent accuracy of the ADP3810 controls the charger and
guarantees the
m
±
1% final battery voltage needed for Li-Ion
charging.
b
1
0
nF
3.3V
1.0
1N4148 1
00
V
V
CTRL
= 1.0V
C
F2
220
0.9
22
m
F
m
F
50
m
F
/
4
50V
TX1**
0.8
0.7
V
OUT
BATTERY
9.1
V
3W
100k
V
C
F1
R4
1.2k
V
1mF
0.6
L
2
2n
F
1N4148
V
CTRL
= 0.5V
1A
AC
120/220V–
0.5
4
7
m
F
R
CS
13V
MURD320
R3
20k
V
*
0.25
V
*
0.4
N
33
0
pF
C
C2
3
30
V
R
C2
0.3
0.2
0.1
V
CTRL
= 0.25V
V
CTRL
= 0.125V
0.
2
m
F
V
CC
OUTPUT
3
00
V
10
V
MAXIMUM V
OUT
= 8.4V
CHARGE CURRENT
0.1A TO 1A
IRFBC30
COMP
10k
V
0.
1
m
F
0.
1
m
F
R
F
C
F
PWM 3845
3.3k
V
1nF
0
2
1
k
V
V
FB
3456789 0
11
I
SENSE
V
REF
V
OUT
3.3k
V
V
CS
V
REF
V
CC
V
SENSE
470pF
1.6
V
CHARGE
CURRENT
CONTROL
VOLTAGE
ADP3810/ADP3811
V
CTRL
0.1
m
F
Figure 2. Current/Voltage Transition
of the ADP3810 CCCV Charger
RT/CT
GND
0.1
m
F
OUT
COMP
GND
* 1% TOLERANCE
** TX1
C
C1
1
m
F
f = 120kHz
L
PR
= 750
3.3k
V
2.2nF
R
C1
10k
V
m
H
L
SEC
= 7.5
m
H
OPTO COUPLER
MOC8103
*For technical data, consult our Web site,
www.analog.com
,
use Faxback (see p. 24).
Figure 3. Complete Off-Line Li-Ion Battery Charger
4
Analog Dialogue 31-2 (1997)
14-Bit Monolithic
ADCs: First to Sample
Faster than 1˚ MSPS
1.25 to 10˚ MSPS pin-compatible
AD924xs enable new applications in
communications and imaging
by Mike Walsh, Larry Singer, and Joe DiPilato
The AD924x family are the industry’s first monolithic 14-bit
analog-to-digital converters (ADCs) to exceed a 1-MHz sample
rate. The three pin-compatible devices in MQFP-44 packages,
AD9240, AD9243, and AD9241 are specified at 10, 3, and 1.25-
MHz clock rates, respectively. With their 12-bit counterparts, the
AD9220/23/21 family, they form a complete set of high-
performance CMOS A/D converter solutions.*
The monolithic single-supply AD924x series of converters at last
offer the benefits of high performance, accompanied by significant
savings of cost, power, and board space. The assembled hybrids
and modules that they will supplant cost many hundreds of dollars,
dissipate watts of power, and are typically packaged in large 24-
pin DIPs; they operate from a minimum of two supplies and are
usually specified for the 0 to 70°C commercial temperature range.
The AD924x family is 5 to 20 times less costly in price and power
than a popular family of competitive hybrids, are smaller, and have
better dynamic specifications. The table lists some of the key
specifications of the AD924x family [SNR (signal-to-noise ratio),
SINAD (signal to noise & distortion), THD (total harmonic
distortion) and SFDR (spurious-free dynamic range)]. The devices
operate from a single 5-V supply and have the low power dissipation
shown.
imaging applications benefit from low power dissipation (heat
generation); the ADC can reside closer to the IR sensor. Yet other
applications for high performance, low power, and low price
include: instrumentation, radar, collision-avoidance systems, test
equipment, signal analysis, and data acquisition.
Like many high speed converters offered by Analog Devices, the
AD924x series is based on a multibit, pipelined architecture, but
it is implemented in low-power switched-capacitor circuitry. Figure
1 shows a block diagram of the complete ADC. A low-noise,
wideband sample-hold amplifier (SHA) with differential outputs
precedes the pipelined core, and accepts single-ended or differential
inputs up to 5˚ V˚ p-p. From the SHA’s output, the signal path is
fully differential. The first pipeline stage converts the 5˚ most
significant bits and amplifies the remainder, or
residue
, for
successive conversions by the next three 4-bit stages. The results
of these partial conversions by the four pipeline stages are then
time-aligned and added (with one bit of overlap) to obtain the final
14-bit result. Each clock cycle produces a new conversion, with
3-cycle latency.
VINA
DAC-AMP1
G1 = 16
DAC-AMP2
G2 = 8
DAC-AMP3
G3 = 8
SHA
VINB
F1
F2
F3
F4
5
4
4
1V
5
4
4
4
VREF
DIGITAL CORRECTION LOGIC
REF
14
DOUT
SENSE
Figure 1. 14-bit pipelined ADC architecture.
The converter’s overall DC accuracy (INL, DNL) largely depends
on the accuracy of the first pipeline stage, which is limited by
capacitor mismatch. By converting 5 bits in the first pipeline stage,
the effects of capacitor mismatch are sufficiently suppressed to
achieve 14-bit accuracy without the need for on-chip calibration.
Integral and differential nonlinearity are typically ± 2.5 and
±
0.6˚ LSB, respectively.
The dynamic and noise performance of the A/D are largely
determined by performance of the input SHA, which was carefully
optimized to provide low noise and distortion over a moderately
wide bandwidth. Typical input-referred noise is 0.36˚ LSB, or
110˚
AD9240 AD9243 AD9241
Update rate (MSPS) 10 3 1.25
AIN frequency (kHz) 500 500 500
SNR (dB typ/min) 78.5/76 80/77 79/75.5
SINAD (dB typ/min) 77.5/75 79/76 78/74.5
THD (dB typ/max) –85/–77 –87/–80 –88/–77.5
SFDR (dB typ) 90 91 88
Power dissipation (W max) 0.33 0.145 0.085
Price ($US, 100s) $74.95 $49.95 $21.50
Their high performance, low power, and low price are of particular
relevance in emerging and next-generation consumer applications,
such as communications and imaging. They will be used in cellular
and PCS basestations, ADSL/HDSL modems, flatbed and drum
document scanners, film and x-ray scanners, infra-red and medical
imagers.
For
communications
, wide input bandwidth, low distortion & wide
dynamic range, and low power are major attractions. Wide dynamic
range helps to reduce gain requirements in the receiver IF strip.
High input bandwidth allows the AD924x family to be used in
undersampling applications to perform IF to baseband down-
conversion/mix-down. For
imaging
, their low noise, 14-bit no-
missing code, and SNR performance are key. In addition, infra-red
V˚ rms. Figure 2 compares typical S/(N+D) and total
harmonic distortion (THD) as a function of input frequency for
the three devices at their specified sampling rates. These plots
demonstrate superior dynamic performance well beyond the
devices’ respective Nyquist frequencies.
m
85
–45
SIGNAL-TO-NOISE
PLUS DISTORTION
AD9240
80
–50
75
–55
AD9243
70
AD9241
–60
65
–65
60
–70
AD9241
–75
55
AD9243
AD9240
TOTAL HARMONIC
DISTORTION
50
–80
–85
–90
45
40
0.01
0.1
1
10
100
FREQUENCY – MHz
Figure 2. SINAD and THD vs. Signal Frequency
The on-chip bandgap voltage reference can be pin-strapped to
1˚ V or 2.5˚ V, or set for any voltage in between using an external
resistor divider. Optionally, an external voltage reference may be
used. The AD924x family, packaged in a 44-pin MQFP, operates
over the –40 to +85°C extended industrial temperature range.
b
*For technical data, consult our Web site,
www.analog.com
, use Faxback (see
p. 24).
Analog Dialogue 31-2 (1997)
5
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